Index error correction for flux valve heading repeater system

ABSTRACT

A flux valve compass heading repeater system is provided with a compensating system which, when connected to a three-legged flux valve, provides fully compensated, three-wire output signals of the synchro data transmitter type for direct use in apparatus requiring precision three-wire heading data. The compensating system includes control circuits for generating sine and cosine components of magnetic heading and for compensating them for typical compass errors such as those induced by changes in operating latitude and two cycle and index errors. Latitude compensation is accomplished by a novel proportional automatic gain control; two cycle cardinal heading error compensation is accomplished by a compensation circuit having only a single manual control, while index error compensation is similarly accomplished by a compensation circuit requiring only a single manual control.

CROSS REFERENCE TO RELATED APPLICATIONS

The present application is directly related to copending applications:

Ser. No. 528,760: "Latitude Compensator for Flux Valve Heading RepeaterSystem," and

Ser. No. 528,758: "Two Cycle Compensator for Flux Valve Heading RepeaterSystem."

These applications were filed concurrently in the name of J. R. Erspamerand include substantially identical disclosures.

BACKGROUND OF THE INVENTION Field of the Invention

The invention pertains to means for the compensation of undesirableerrors in the signal characteristic of three-wire data repeater systemsand more particularly relates to the correction of adverse effects whichcause undesired variations in the outputs of flux valve compass datarepeater systems, including errors due to variation in the horizontalcomponent of the earth's field, index angle errors, and cardinal andintercardinal heading errors.

Description of the Prior Art

When navigating at high latitudes with flux valve magnetic compasssystems, difficulty is experienced because of the decreasing strength ofthe horizontal component of the earth's magnetic field as latitudeincreases. The usual flux valve providing the input to the magneticcompass system normally senses only the horizontal component of theearth's field. At high latitudes the consequence is that the strength ofthe sensed horizontal component of the earth's field is proportionallylessened, and the compass system experiences decreasing sensitivity,resulting in heading information of diminished accuracy.

Prior art systems have sought to solve this compensation problem ofproviding an input to the compass data repeater substantiallyindependent of variations in the strength of the horizontal component ofthe earth's magnetic field by controlling the gains of amplifiers or theeffective values of impedances in the separate channels of the datatransmitter system in a relatively complex manner, but generally ininverse relation to the signal strength as measured at the flux valveitself. Examples of such arrangements are described by D. A. Espen inthe U.S. Pat. No. 3,548,284 for "Synchro Data Transmission ApparatusHaving Discrete Gain Changing to Compensate for Undesirable SignalGradient Variations" issued Dec. 15, 1970, and by J. R. Erspamer and G.W. Snyder in the U.S. Pat. No. 3,646,537 for an "Automatic Gain Controlfor an Electromechanical Transducer", issued Feb. 29, 1972, both patentsbeing assigned to the Sperry Rand Corporation. While these concepts havebeen useful in providing adequate magnetic field compensation under manycircumstances, the compensating signals compensate only for variation inthe horizontal magnetic field components, and generally do notadditionally correct fully for gain changes caused by componentvariations or due to temperature or power supply voltage drifts or tocomponent aging. Further, the characteristics of the individual gaincontrol elements of the individual channels of the data system may varywithout proper corrective relative adjustments, resulting in thegeneration of two-cycle transmission errors in gain control stages.

The improved system disclosed by J. R. Erspamer and G. W. Snyder for a"Multiplexed Gain Control for Synchro Data Transmission System" in theU.S. Pat. No. 3,784,753, issued Jan. 8, 1974, sought more fully toovercome these prior art defects by a relatively complex and expensivecorrection circuit. Though it generally overcame such defects, it wasfound that some undesirable two cycle error could be generated in itsrelatively complex automatic gain control stage, and that a simple waywas needed for identically changing the gains of channels of the datatransmission system, but retaining the advantages of the concept of U.S.Pat. No. 3,784,753.

Prior art systems have additionally sought to provide correction for theindex angle error in compass data transmission systems by use ofnetworks including precision differential synchros or ganged dualpotentiometers which must track each other with high precision if theyare not themselves to introduce errors. According to the presentinvention, the expense of obtaining such selected precision synchros orpotentiometers is desirably eliminated. Cardinal heading error wassimilarly corrected in prior compass data transmission systems by usinga network with precision ganged dual potentiometers of similar qualityand it is found increasingly desirable to substitute simpler and lessexpensive networks permitting use of only a single adjustment controlfor each of these correction purposes, at the same time retaining a highdegree of precision.

SUMMARY OF THE INVENTION

The present invention provides means for the correction of undesirablechanges in signal amplitudes in multiple channel flux valve datarepeater systems partly by the employment of a simple common automaticgain control in a circuit configuration which not only compensates formagnetic field strength changes, but also corrects for the effects ofother error sources without introducing the errors of prior art systems.The novel control system monitors the data repeater signals near theinputs to the utilization device, rather than merely at the outputs ofthe flux valve. By monitoring the inputs at the utilization device andby using the data repeater excitation voltage as a switching reference,the gain control, being part of a closed feed back loop, additionallycompensates for gain changes caused by variations of componentparameters and for other effects without itself introducing new errors.According to a primary aspect of the present invention, electricallycross coupled network means provides correction in the sine and cosinechannels of the flux valve data transmission system for any index errorangle correction by the adjustment of only a single control. A similararrangement, again requiring only one adjustment, is employed forcorrection of the two cycle cardinal heading error. In a modification ofthe system, the latter two compensations are applied in the form ofdirect current signals in the correct ratios coupled directly into theflux valve inductive windings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1A and 1B illustrate, partly in block diagram form, the principalelements of the invention and their electrical interconnections.

FIG. 2 is a portion of FIGS. 1A and 1B showing details of the novelautomatic gain control circuit.

FIG. 3 is a detailed circuit diagram of the novel index error anglecompensator of FIG. 1B.

FIG. 4 is a detailed circuit diagram of the novel two cycle compensatoralso of FIG. 1B.

FIG. 5 is a block diagram of an embodiment alternative to that of FIGS.1A and 1B.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

In FIGS. 1A and 1B, the novel compensated compass system includes amagnetic azimuth detector or flux valve 11 which may be of the generaltype disclosed in the M. C. Depp U.S. Pat. No. 2,852,859 for a "FluxValve Compensating System," issued Sept. 23, 1958 and assigned to SperryRand Corporation. Other details of such flux valve devices are disclosedin the D. J. Kesselring U.S. Pat. No. 3,573,610, issued Apr. 6, 1971, inthe D. J. Kesselring et al. U.S. Pat. No. 3,641,679, issued Feb. 15,1974, and in the U.S. Pat. application Ser. No. 380,523 for "A FluxValve Apparatus for Sensing Both Horizontal and Vertical Components ofan Ambient Magnetic Field," filed July 18, 1973, issued as U.S. Pat.No., Mar. 25, 1975 and assigned to Sperry Rand Corporation. Flux valve11 is excited by alternating current source 2, which may be aconventional 400 Hz. oscillator or signal generator and which is coupledto excitation winding 12 of the flux valve 11.

As disclosed in the aforementioned Depp and Kesselring patents, fluxvalve 11 has three wye-connected inductive windings 13, 14, and 15 on acorresponding wye-shaped core, the winding legs meeting at a commongrounded terminal F. The terminals of windings 13, 14, and 15 oppositeterminal F are respectively labelled A, B, and C. Terminals A, B, and Cmay, if desired, be supplied with single cycle compensation signals froma single cycle compensator (not shown) of the general type shown in theaforementioned U.S. Pat. No. 2,852,859.

Terminal A of flux valve 11 is connected via a blocking capacitor 16 toone winding 20 of a Scott tee transformer 21, while terminals B and Care connected via respective blocking capacitors 17 and 18 to therespective ends of a second input winding 22 of Scott tee transformer21. Winding 22 has a center tap connected to the other end of winding20.

As is well known, the signal outputs of windings 13, 14, and 15 have afrequency double that applied to excitation winding 12. The frequencydoubled cosine output of winding 23 of transformer 21 and its frequencydoubled sine output in winding 27 are connected to current servo loop31. Additionally supplied to current servo loop 31 via lead 29a is theoutput of frequency doubler 29. Since frequency doubler 29 is excited bygenerator 2, its output on lead 29a will have an 800 Hz. frequency andserves as a reference signal source for servo 31.

As described in detail in the D. H. Baker, F. H. Kallio U.S. Pat. No.3,678,593 for a "Compass System and Components Therefor Having AutomaticField Cancellation", issued July 25, 1972 to Sperry Rand Corporation,current servo 31 supplies outputs on leads 32 and 33 which are directcurrent signals respectively proportional in amplitude to the sine andcosine of craft magnetic heading (H_(m) sin ψ and H_(m) cos ψ).Accordingly, the horizontal components of the earth's magnetic fieldsensed by the flux value windings 13, 14, and 15 are resolved into sineand cosine component values that are then converted by current servo 31into proportional direct currents on leads 32 and 33. As taught in theaforementioned Baker et al. patent, these direct current components arefed back via leads 10 and 10a into windings 13 and 15 of flux valve 11,which currents tend to cancel the earth's magnetic field therein. Thefeed back arrangement and its many advantages are discussed in detail inthe aforementioned U.S. Pat. No. 3,678,593, including closed loopoperation affording high accuracy outputs in the form of analog directcurrent outputs proportional to the sine and cosine of craft magneticheading.

Accordingly, the 800 Hz., three-wire magnetic azimuth informationderived by the horizontal magnetic field detector or flux valve 11 isconverted to direct current signals proportional to the sine and cosineof craft heading by the cooperation of Scott tee transformer 21 andcurrent servo 31. The magnitudes of the outputs on leads 32 and 33 arethus a function of craft magnetic azimuth or heading and the intensityof the horizontal component of the earth's magnetic field. The variationin the magnitude of the sine and cosine outputs caused by any change inmagnetic field strength H_(m) affects only the output gradient (voltsper azimuth degree) and does not change the trigonometric relationshipof the input magnetic heading angle ψ and the output voltages of currentservo 31, which may therefore be expressed as follows:

    V.sub.32 = K.sub.1 sin ψ                               (1)

and

    V.sub.33 = K.sub.1 cos ψ                               (2)

where K₁ allows for the gain of current servo 31 and has dimensions ofvolts per oersted.

The signals V₃₂ and V₃₃ on leads 32 and 33 serve as two inputs to theautomatic gain control circuit 34, which circuit also receives certainfed back signals on leads 56 and 57. As will be further discussed, thefed back signals arise at the outputs of buffer amplifiers 52 and 53after the outputs of automatic gain control 34 are processed at least bydual channel modulator 45. To understand the operation of the gaincontrol circuit 34, the presence of the index error compensator 37 andthe two-cycle compensator 48 may be ignored for the moment as a matterof convenience.

The final output of the compass system supplied by leads 61, 62, and 63to an aircraft navigation system or other utilization device 64 isusually required to be useful in a three-wire synchro data transmittersystem and to consist of proportional voltages between pairs of suchleads, as between leads 61 and 62, 62 and 63, 63 and 61. These maynominally be 11.8 volts, for example, and must be maintained at aconstant gradient in the interest meeting required compass accuracy overa wide range of horizontal magnetic field strengths H_(m). Because theoutput of flux valve 11, and therefore the output of current servo 31,has a gradient which is directly proportional in magnitude to thehorizontal magnetic field strength which, of course, varies withlatitude, the automatic gain control stage 34 is required to hold thesystem output signals at leads 61, 62, and 63 at the desired nominal11.8 volt leg-to-leg constant gradient.

For this purpose, the direct current outputs on leads 35 and 36 of gaincontrol 34 are supplied to the conventional dual channel modulator 45,each of the two individual channels of which are supplied by lead 2awith the 400 Hz. reference signal output of generator 2. The directcurrent signals on leads 35 and 36 are modulated by the 400 Hz.alternating current signal in the conventional manner so that 400 Hz.signals appear on leads 46 and 47, proportional respectively to the sineand cosine of the magnetic heading of the craft. After individuallyseparate supply to buffer amplifiers 52, 53, equally amplified versionsof these signals appear on leads 54, 55 to which the feed back leads 56and 57 are respectively connected.

The automatic gain control 34, shown in greater detail in FIG. 2,monitors the gradient at the output leads 54, 55 of buffer amplifiers52, 53, respectively, compares the result to a reference voltage level,then varies the system gain accordingly by control of the gain ofautomatic gain control circuit 34. If the gradient at the outputs ofbuffer amplifiers 52, 53 of FIG. 1A is less than a predetermined level,the voltage gain of circuit 34 is increased to bring the output of thebuffer amplifiers 52, 53 up to the proper level. The output of thebuffer amplifiers 52, 53 and the voltage gradient is similarlycontrolled. The signal levels at output leads 35, 36 of the automaticgain control 34 are ultimately passed through the output Scott teetransformer 60. The outputs of transformer 60 are therefore fullyindependent of any earth's magnetic field strength variation. Thus:

    V.sub.35 = K.sub.2 sin ψ                               (3)

and:

    V.sub.36 = K.sub.2 cos ψ                               (4)

where K₂ is a new proportionality constant.

Automatic gain control circuit 34 is designed to prevent theintroduction of any stand-off or unbalance between transmissionchannels, resulting in cyclic errors, into the craft heading outputdata. The individual gains of the sine and cosine channels are nowidentically controlled and there are no off-set voltages induced intothe direct current signals representing sine and cosine of craftmagnetic heading. As is seen in more detail in FIG. 2, the directcurrent signals at leads 32 and 33 are, as before, provided by thecooperative action of flux valve 11, Scott tee transformer 21, andcurrent servo 31, and are respectively proportional in amplitude to sinψ and cos ψ. Output lead 32 is coupled in series through resistor 75,junction 76, resistor 77 and input lead 35 to one channel of the dualmodulator 45. At input lead 35 is a capacitor 78 coupled to ground andforming a low pass filter with resistor 77. Likewise, the second outputlead 33 is coupled in series through resistor 79, junction 80, resistor81, and input lead 36 to a second channel of dual modulator 45. At inputlead 36 is a capacitor 82 forming a low pass filter with resistor 81.Switching or chopper transistors 83 and 84 are respectively coupled toground from junctions 76 and 80 and control current flow through theiremitter and collector electrodes in accordance with their respectivebase voltages.

The direct current signals on leads 32 and 33 are chopped by transistors83 and 84, respectively and, after smoothing by low pass filters 77-78and 81-82, form direct currents that are individually modulated in dualchannel modulator 45 by the 400 Hz. reference signal on lead 2a. Thesedual channel output voltages are directed by Scott tee transformer 60 asthree-wire synchro data to a navigation system or other utilizationdevice 64.

For purposes of controlling the automatic gain control circuit 34, thesame 400 Hz. modulated output currents on leads 54 and 55 arerespectively coupled by leads 56 and 57 to a constant amplitude,variable phase circuit comprising resistor 95 and capacitor 96 coupledin series with leads 56, 57 at junction 97. Circuit 95-96 is of thegeneral kind discussed in the D. A. Espen U.S. Pat. No. 3,548,284,entitled "Synchro Data Transmission Apparatus Having Discrete GainChanging to Compensate for Undesirable Signal Gradient Variation",issued Dec. 15, 1970 and in the D. A. Espen U.S. Pat. No. 3,617,863,entitled "Constant Amplitude Variable Phase Circuit," issued Nov. 2,1971, both patents being assigned to Sperry Rand. The constantamplitude, variable phase signal found at junction 97 is rectified bydiode 94 and appears as a variable unipolar voltage at one input of aconventional integrating operational amplifier 92 having its outputcoupled by capacitor 91 to its same input. To the second input ofamplifier 92 is coupled through resistor 93 a stable positiveunidirectional reference voltage from a suitable source (not shown)connected to terminal 98. As shown in the drawing, amplifier 92 and itsassociated circuit act as conventional comparator means for in effectcomparing the output gradient on leads 54, 55 with the fixed levelvoltage at terminal 98, yielding an integrated output as a function ofthe difference of the two voltage levels at leads 97 and 98.

The positive signal at the output of device 92 is coupled throughresistor 88 to one input of amplifier 87, to the other input of which issupplied at terminal 90a and through resistor 89 the 400 Hz. excitationsignal from generator 2. Under control of its varying amplitude directcurrent and constant amplitude alternating input currents, circuit 87acts as a conventional variable pulse-width generator for supplying a400 Hz. variable pulse-width signal at junction 74.

The variable pulse-width signal is coupled in parallel from junction 74through the respective resistors 85, 86 to the base electrodes ofchopper transistors 83 and 84 to control the relation of the conductionto non-conduction times of these switching transistors. The transistors83, 84 are synchronously conducting at the same time and then are bothnon-conducting for a controlled period of time depending upon the pulsewidth of the output of amplifier 87. As the non-conducting part of thecycle is increased in time duration, the total current per cycle passingfrom lead 32 to lead 35, for example, is increased. In other words,proportionately less of the current available on lead 32 is dumped toground. In this manner, the voltage between leads 54, 55 is madeindependent of any amplitude variations in the total flux valve data aswell as amplitude variations resulting from other disturbing factors inthe signal channels between current servo 31 and buffer amplifiers 52and 53. Accordingly, the three-wire output supplied to utilizationdevice 64 of FIG. 1 by transformer 60 is maintained nominally constantfrom leg to leg, such as at 11.8 volts.

In the complete system as illustrated in FIG. 1, the outputs V₃₅ and V₃₆of the automatic gain control 34 on leads 35 and 36 may be firstprocessed by the novel index error angle compensator 37 prior to 400 Hz.modulation. For this purpose, the compensation circuit of FIG. 3 isemployed. The index error angle compensated by circuit 37 is presentbecause of the normal difficulty of achieving perfect alignment betweenthe aircraft fore-aft axis and the effective electrical fore-aft axis ofthe flux value 11. Accordingly, index angle error compensator 37 isprovided to permit a manual correction to be made after systeminstallation by performing, in essence, the same function as might beprovided by a relatively expensive servo differential which some priorart systems have employed. However, since installation accuracies areusually within ±10°, the compensation function may be accuratelyperformed by the relatively inexpensive circuit of FIG. 3 wherein only asingle potentiometer shaft need be adjusted. It will be apparent thatthe correction is made by the novel compensator herein disclosed to thevalue of angle ψ when it is still in the trigonometric form of sin ψ andcos ψ data.

Accordingly, the apparatus of FIG. 3 accepts two inputs K₂ sin ψ and K₂cos ψ and internally generates two values - K₂ β cos ψ and - K₂ β sin ψ.The K₂ sin ψ value and the - K₂ β cos ψ value are added according to thewell known trigonometric identity to form K₂ sin (ψ + β) where ψ' = ψ +β may be used to represent a corrected value of ψ. The K₂ cos ψ valueand the -K₂ β sin ψ values are similarly added to form K₂ cos (ψ + β).In accord with the teachings of the present invention, the β terms mustbe identical in both the sine and cosine output channels to effectprecise compensation; the same source for the β term is used in the twochannels of the circuit.

In greater detail, the circuit of FIG. 3 has, in operation, negativevalued direct voltages representing K₂ sin ψ and K₂ cos ψ as respectiveinputs on leads 35, 36, and these are respectively supplied directly toinputs of conventional unity gain output amplifiers 145 and 155 at theright side of the figure. The same two direct negative voltages are usedin the remaining or major part of the circuit to produce compensatingvoltages also for insertion into amplifiers 145, 155. For the latterpurpose, the -sin ψ term on lead 35 is coupled through a conventionalinverting amplifier 103 to the switching transistor 107. Amplifier 103has its output terminal 104 coupled through a resistor 102 to its inputterminal 101 and additionally has a second input terminal coupled toground through resistor 105. The -cos ψ term on lead 36 is coupleddirectly to switching transistor 109. Transistors 107 and 109 are madealternately fully conducting and fully non-conducting so that, first,the output of amplifier 103 appears on lead 108a and then, the signalpassed by switching transistor 109 appears on lead 108 b. Since both ofthe leads 108a and 108b are coupled to the adjustable contact 113a ofpotentiometer 113, it is seen that the signals alternately passed byswitching transistors 107 and 109 are alternately applied to contact113a for time-sharing purposes in the shared amplifier 120.

The switching transistors 107 and 109 are made alternately conductingunder control of a sine wave signal appearing on lead 2b; this signal isconveniently obtained from the 400 Hz. generator 2 of FIG. 1A, thoughother regular stable-frequency signals may alternatively be employed. Inpractice, the 400 Hz. cycle signal on lead 2b is applied by lead 106 tocontrol the conduction of transistor 107. So that time sharing may beemployed, the signal on lead 2b is coupled via lead 111, the 180° phaseshifter 112, and lead 110 to control the operation of switchingtransistor 109.

In this manner, the signals on leads 35 and 36 are alternately suppliedat the selected contact point of potentiometer 113, the latter havingits opposed terminals 113b and 113c coupled to inputs of operationalamplifier 120. The output terminal 121 of amplifier 120 is coupled toits input at terminal 113b via resistor 115, and terminal 113c isconnected through resistor 114 to ground in conventional fashion. Theinput of amplifier 120 is thus time shared and its output on terminal121 is supplied to a second pair of switching transistors 122, 123,these transistors being arranged for controlling the series signal flowthrough the respective resistors 126, 127 to amplifiers 128, 129. Theeffective gain of amplifier 120 is changed according to the setting ofthe single control 37a, which control is manually set in accordance withthe known magnitude of the index error determined as a result ofconventional ground swinging operations.

Conductivity of switching transistor 122 occurs simultaneously with theconductivity of switching transistors 109. In like manner, conductivityof switching transistor 123 is made simultaneous with the periods ofconductivity of switching transistor 107. This operation is accomplishedby controlling the conductivity of switching transistor 123 according tothe signal on lead 2b when supplied directly to switching transistor 123via lead 125. The desired synchronous operation of switching transistor122 is accomplished by providing the 180° phase shifted signal fromcircuit 112 via lead 124 to transistor 122. In this manner, bothchannels of the circuit time share the use of the common amplifier 120,ensuring that identical corrections are applied to the two channels;i.e., that the amount of the sine term added to the cosine term isidentical to the amount of the cosine term subtracted in the sinechannel. It is further observed that adjustment of the single control37a allows adjustment of potentiometer 113 so that both channels areidentically set in accord with the magnitude of the index error.

The time shared currents alternately flowing through switchingtransistors 122, 123 are alternately supplied to the conventional unitygain amplifiers 128, 129, and the respective outputs on the terminals132, 133 flow through resistors 141, 150 to the same respective inputterminals of amplifiers 145, 155, as are connected to the respectiveleads 35, 36. The outputs of amplifiers 145 and 155 may be smoothed bythe action of appropriate low pass filters so as to remove any 400 Hz.modulation from the outputs appearing in the respective output leads 38,39. In the embodiment illustrated, the filters are placed at the inputsof amplifiers 128 and 129 and comprise resistors 126, 127 and capacitors128a, 129a, respectively.

The mathematical relation expressing the index error as a function ofthe sine and cosine of magnetic heading is: ##EQU1## where: ψ' = thecompensated output,

ψ = the uncorrected input, and

β = the tangent of the index error.

Thus, by adjusting the gain of the time shared amplifier 120 inaccordance with the value of β, expression (5) is satisfied as follows.The output of amplifier 128 is:

    V.sub.132 = -K.sub.2 β cos ψ                      (6)

when transistors 109 and 122 are conducting, and the output of amplifier129 is:

    V.sub.133 = -K.sub.2 β sin ψ                      (7)

when transistors 107 and 123 are conducting. The addition at amplifier145 produces on output:

    V.sub.38 = K.sub.2 (sin ψ - β cos ψ)          (8)

while the addition at amplifier 155 produces an output:

    V.sub.39 = K.sub.2 (cos ψ + β sin ψ)          (9)

or:

    V.sub.38 = K.sub.2 sin ψ'                              (10)

and:

    V.sub.39 = K.sub.2 cos ψ'                              (11)

These direct current signals are ready for conversion in dual channelmodulator 45, providing as they do direct current signals in terms of ψ'containing the desired index angle error compensation as set forth inequation (5). Thus, the dual channel modulator 45 supplies on its outputleads 50, 51 400 Hz. signals whose amplitudes are:

    V.sub.46 = K.sub.3 sin ψ'                              (12)

    V.sub.47 = K.sub.3 cos ψ'                              (13)

and these signals serve as inputs to the two cycle error compensator 48.

Two cycle error in a magnetic field sensor, including the flux valvetype of sensor disclosed herein, is induced by the presence of a softiron mass or masses in the vicinity of the flux valve which tends todistort the earth's ambient magnetic field thereat. As the name implies,the error is a sinusoidal error and has two complete cycles within 360°of azimuth rotation of the craft. In general, the average location ofthe soft iron mass relative to the flux valve determines the directionof its effective vector. For convenience, the two cycle compensation isaccomplished by effectively breaking down the total vector intoorthogonal components, one termed the cardinal two cycle error componentand the other the intercardinal error component. The cardinal two cycleerror has extremum values at heading angle values 0°, 90°, 180°, and270°. Intercardinal two cycle error, on the other hand, has extremumvalues at 45°, 135°, 225°, and 315° azimuth values. In the invention ofFIG. 4, the latter error is readily corrected by placing an adjustableseries resistor 199 in the feed back path of a.c. amplifier 200 that isexcited by lead 46. Adjustment of control 48b in accord with data takenduring installation compass swings then corrects the output at lead 50in the appropriate manner. The intercardinal two cycle heading error iscompensated by changing the gain balance between the sine and cosinechannels supplying outputs on the respective leads 50, 51.

Correction of the cardinal heading error is accomplished by the simplecircuit of FIG. 4, using a single adjustment 48a and a common circuitstage in a manner minimizing error sources and characterized bysimplicity. The amount of adjustment of control 48a is also determinedby the installation ground swinging process. The K₂ sin ψ' signal onlead 46 is supplied via lead 175 through resistors 177 and 180 to therespective inputs of differential operational amplifier 188. The valueK₂ cos ψ' from lead 47 is added through resistor 178 at terminal 183 tothe K₂ sin ψ' term; similarly, the term K₂ sin ψ' from lead 46 is addedthrough resistor 180 at terminal 185 to the K₂ cos ψ' term. Amplifier188 has a resistor 187 coupled between its output 189 and the inputterminal 183 in conventional fashion. A variable resistor 186 with anadjustable control 48a is coupled between terminal 185 and ground. Thevariable resistor 186 constitutes the single cardinal heading erroradjustment, its variation affecting the effective gain γ of amplifier188. According to the setting of control 48a, a compensating voltageappears at the output 189 of amplifier 188:

    V.sub.189 = - γ (sin ψ' + cos ψ')            (14)

The signal value V₁₈₉ is coupled at junction 190 to branching leads forsupplying this signal through resistors 192, 193 to the inputs ofrespective amplifiers 200 and 201. As noted previously, amplifier 200has a variable resistor 199 coupled between its output 203 and its inputlead 195. The other input to amplifier 200 is connected through resistor196 to ground. A further amplifier 201 is supplied with the signal K₂cos ψ' from lead 47 through resistor 194 and is similarly provided witha resistor 202 connecting its output 204 to its input lead 197. Itsimilarly employs a resistor 198 coupled between a second input andground. Amplifiers 200 and 201, through the respective connectors 195and 197, serve as adding and inverting circuits so that the K₂ sin ψ'term on lead 46 has added to it the correction term appearing atterminal 190 and the summation is found on output lead 50. In a similarmanner, the K₂ cos ψ' signal supplied on lead 47 is added to thecompensating signal on junction 190 by amplifier 201 and its associatedcircuit, an inverted signal being generated on output lead 51. In thismanner, the voltage V₅₀ is:

    V.sub.50 = K.sub.4 [sin ψ' + γ (sin ψ' + cos ψ')](15)

and that on output lead 51 is:

    V.sub.51 = K.sub.4 [cos ψ' + γ (sin ψ' + cos ψ')](16)

In equations (15) and (16), the new value K₄ may include the effect ofthe adjustment of resistor 199. Thus, the voltage on output lead 50 issin ψ" and the voltage on output lead 51 is cos ψ", where ψ" representsψ' corrected both for cardinal and intercardinal heading errors. Fromequations (15) and (16), it is evident that the value of the correctedangle ψ" is expressed by the following equation: ##EQU2## ψ" being thefinal output heading value components corrected for cardinal andintercardinal two cycle errors. It is seen that amplifier 188, theeffective gain γ of which is controlled by the setting of the variablepotentiometer 186, cooperates in the circuit in generating the functionγ (sin ψ' + cos ψ'), and this function is added in the sine and cosinechannels by the respective action of amplifiers 200 and 201 and theirassociated circuits. It is observed that correction of the cardinalheading two cycle error is accomplished by manual operation of a singleadjustment. Furthermore, the single stage associated with amplifier 188minimizes potential error sources and aids in simplifying the adjustmentprocedure.

It will be understood that the invention may be employed in alternativeforms and that the compass system of FIGS. 1A and 1B may be modifiedwithin the scope of the claims appended hereto, for example, asillustrated in FIG. 5. In the embodiment of FIGS. 1A and 1B, the indexcompensation and two cycle error compensation signals are generated fromthe sine and cosine outputs of the current servo 31 and are re-applieddownstream in the two channels to be summed with their original oruncompensated values. In the modification illustrated in FIG. 5, thesine and cosine outputs of the current servo 31 are used in essentiallythe same manner to generate the compensating signals as direct currentsignals; however, the summing of these signals with the original data isaccomplished directly at the flux valve 11 by feeding back thecompensating signals as direct currents into the flux valve legsthemselves so as to, in effect, compensate the output of the flux valveitself broadly in accordance with the concept of the above referencedDepp U.S. Pat. No. 2,852,859.

Referring now to FIG. 5, similar reference numerals are used todesignate elements corresponding to those found in FIGS. 1A through 4;elements not found in the latter figures are identified by referencenumerals in the three hundreds. It will be seen that the embodiment ofFIG. 5, like that of FIGS. 1A and 1B, employs in serial array areference signal generator 2, a flux valve 11, blocking capacitors 16,17, and 18, an input Scott tee transformer 21, a current servo 31, anautomatic gain control 34, a dual channel modulator 45, power amplifiers52 and 53, an output Scott tee transformer 60, and a utilization device64. In a manner generally similar to that employed in FIG. 1A withrespect to current servo leads 10 and 10a, the respective correctioncurrents are fed to summation points 320, 321 of FIG. 5 so that theyflow through the respective legs of flux valve 11 to ground, beingblocked from flowing into the Scott tee transformer 21 by capacitors 16,17, and 18.

In the lower portion of FIG. 5, the index compensation circuit 37 isschematically illustrated. The similarity with the correspondingstructure of FIG. 3 will be immediately apparent and the simplificationof the illustration correspondingly apparent. Thus, the sin ψ and cos ψdirect current signal outputs of the current servo 31 on leads 32, 33,respectively, are alternately applied after one is inverted by inverter340 to the input of a variable gain amplifier 120a through switch means300 corresponding to transistor switches 107, 109 of FIG. 3. The gain ofamplifier 120a is illustrated schematically as being controlled by anadjustment knob 37a corresponding generally to the gain control ofamplifier 120 of FIG. 3 by knob 37a and potentiometer 113 in accordancewith the value β. The output of amplifier 120a is similarly alternatelyswitched to two branch leads as in FIG. 3 by means of switch 301corresponding generally to transistor switches 122, 123 of FIG. 3. Thecontrol of switches 300 and 301 of FIG. 5 is the same as that of FIG. 3,but is illustrated for convenience schematically in FIG. 5 by switchcontrol means 302 controlled, for example, by the 400 Hz. source 2. InFIG. 3, the outputs of switches 122 and 123 are applied to two branchingcircuits including amplifiers 128 and 129 for modifying or summing withthe original sin ψ and cos ψ direct current outputs of the current servo31 through amplifiers 145 and 155. On the other hand, two outputbranches of switch 301 of FIG. 5 are applied correspondingly to controldirect current flow for supplying compensating currents in the properratio to the 120° spaced inductor coils 13, 14, and 15 of flux valve 11for effective summing with the original sources of the sin ψ and cos ψsignals of the current servo 31. In FIG. 5, these direct current ratiosare determined by the selected resistors 303, 304, and 305, in theratios indicated. The currents from resistors 304 and 305 are applied toflux valve winding 15, while that from resistor 303 is supplied to fluxvalve winding 13. If desired, the resistor-capacitor circuits 306 and307 may be used to reduce transient effects of switches 300, 301.

Thus, as in FIG. 3, the apparatus of FIG. 5 serves to provide indexerror compensation through the time sharing of a single amplifier 120abetween the sin ψ and cos ψ channels by alternate operation of theswitches 300, 301, the gain of the amplifier 120a being controlled by asingle control element 37a in accordance with the magnitude of theerror. Such operation insures that the amount of sin ψ current suppliedto the flux valve legs 13, 14, 15 and contributing to the cos ψ outputchannel of current servo 31 is identical to the amount of cos ψ currentsubtracted from the flux valve legs 13, 14, 15 and contributing to thesin ψ output channel of current servo 31.

A modification of the cardinal and intercardinal two cycle errorcompensator of the heading repeater system of FIG. 4 is shown in FIG. 5.Again, the significant feature of the embodiment resides in the mannerin which the compensating signals are summed with the primary signals;i.e., at the flux valve 11 rather than at the output of the currentservo 31. In FIG. 5, for the cardinal two-cycle error compensation, thesin ψ and cos ψ direct current outputs of current servo 31 on leads 32and 33 are summed together in a summing circuit, schematicallyillustrated at 310, prior to supply to the input of variable gainamplifier 188a. The summing circuit 310 of FIG. 5 corresponds to thesumming network 177-180 of FIG. 4, while the gain of the amplifier 188ais illustrated schematically as being varied by the adjustment of knob48a in accordance with the magnitude γ corresponding to the tangent ofthe desired correction. As in FIG. 4, the output of amplifier 188a iscoupled through branching leads and the respective cooperating resistors311 and 312 to summation circuits 320 and 321. Instead of being addedback into the uncompensated sin ψ and cos ψ channels at amplifiers 200and 201 in FIG. 4, the compensation currents are employed in thestrengths indicated in the drawing of FIG. 5 so as to be fed directlyinto the winding legs 13 and 15 of flux valve 11. The currents areeffectively summed with the flux valve winding outputs which contributeto the sin ψ and cos ψ signal outputs of the current servo 31.

The intercardinal two cycle error compensating signal is similarlyapplied to the flux valve windings. The direct current sin ψ signaloutput of current servo 31 is applied to variable gain amplifier 199a,the gain of which is varied by knob 48b in accordance with the magnitudeof the required corrections, amplifier 199a of FIG. 5 corresponding tothe variable impedence 199 and amplifier 200 of FIG. 4. The output ofamplifier 199a is modified by resistors 313 and 314 in accord with theratios indicated in FIG. 5 for application to the respective summationelements 320 and 321 and thus to the windings 13 and 15 of flux valve11, so that the intercardinal two cycle correction signal is effectivelysummed with the flux valve winding outputs contributing to the outputsof the current servo 31.

Thus, in the modification of FIG. 5, the index and two cycle errorcompensation signals are generated from the sin ψ and cos ψ directcurrent outputs of the current servo 31 and are then fed back into theappropriate flux valve inductor windings in the required ratios so thatthe flux valve output supplied to the current servo 31 is compensated.It will be noted in the FIG. 5 embodiment that the feedback compensationsignals are generated from the current servo outputs prior to thelatitude compensation automatic gain stage 34. This is desirable becausethe compensating direct current signals supplied to the valve windingsare essentially associated with the direction of the magnetic fieldsensed by the inductors and in this sense are not related to thelatitude gain compensation.

While the invention has been described in its preferred embodiments, itis to be understood that the words which have been used are words ofdescription rather than of limitation and that changes within thepurview of the appended claims may be made without departing from thetrue scope and spirit of the invention in its broader aspects.

I claim:
 1. In a magnetic compass data transmission system for navigablecraft having means responsive to the earth's magnetic field forgenerating first and second unidirectional output signals proportionalto sin ψ and to cos ψ and substantially representative of said earth'smagnetic field direction ψ with respect to said craft and substantiallyindependent of the magnitude thereof, the combination for correctingcraft index error angle comprising:first time shared amplifier circuitmeans cyclically and alternately responsive to said signals proportionalto sin ψ and cos ψ,said time shared amplifier circuit means having anadjustable effective gain β representative of said craft index errorangle and providing cyclic and alternate output signals proportional toβ cos ψ and β sin ψ at a single output thereof, second amplifier circuitmeans cyclically responsive for passing said cyclic and alternatingsignals proportional to β cos ψ and β sin ψ to separated first andsecond respective outputs thereof, first summation means responsive tosaid second amplifier circuit means for forming a signal proportional tothe difference between said signals proportional to sin ψ and β cos ψand representing said data transmission system first unidirectionaloutput signal corrected for the effect thereon of craft index errorangle, second summation means responsive to said second amplifiercircuit means for forming a signal proportional to the sum of saidsignals proportional to cos ψ and β sin ψ and representing said datatransmission system second unidirectional output signal corrected forthe effect thereon of craft error angle, and utilization meansresponsive to said first and second summation means.
 2. Apparatus asdescribed in claim 1 wherein said first time shared amplifier circuitmeans comprises:dual input amplifier means coupled across potentiometermeans having adjustable contact means, first current switching means forcyclically conducting and not conducting said signal proportional to sinψ with respect to said contact means under control of a referencealternating signal, and second current switching means for cyclicallyconducting and not conducting said signal proportional to cos ψ withrespect to said contact means under control of said referencealternating signal shifted in phase by 180°.
 3. Apparatus as describedin claim 2 wherein said second amplifier circuit means includes:firstcurrent switching means for cyclically conducting said signalproportional to sin ψ to said second summation means under control ofsaid reference alternating signal, and second current switching meansfor cyclically conducting said signal proportional to cos ψ to saidfirst summation means under control of said reference alternating signalshifted in phase by 180°.
 4. In a magnetic compass system for navigablecraft including a magnetic field detector of the flux valve type mountedin the craft for sensing the direction of the earth's magnetic fieldrelative to the craft, the combination for correcting for any errors inthe orientation of said detector relative to the direction axis of saidaircraft comprising,magnetic field detector means including a pluralityof inductor elements angularly disposed on said craft for sensing thedirection and magnitude of the correspondingly disposed horizontalcomponents of the earth's magnetic field relative to the craft and forproviding a corresponding plurality of alternating signals proportionalthereto, signal processing means coupled with said inductor elements andresponsive to said plurality of alternating signals for providing firstand second direct current signals proportional in sense and magnitude topredetermined functions of said earth's magnetic field direction,amplifier means having an input and an output and means for effectivelycontrolling the gain thereof, means for modifying said first and seconddirect current signals in accordance with the output of said amplifiermeans, input and output switching means at the respective input andoutput of said amplifier means, said input switching means beingconnected to receive said first and second signals and said outputswitching means being connected to supply the output of said amplifiermeans to said modifying means, means alternately and simultaneouslycontrolling said switching means such that when said amplifier means isswitched to receive said first signal its output is switched to supplythe means for modifying said second signal and vice versa whereby toinsure that the amount of said first signal modifying the second signalis identical to the amount of said second signal modifying said firstsignal, and, means for controlling said amplifier gain control means inaccordance with said orientation error.
 5. The combination set forth inclaim 4 wherein said first and second signals are proportional to thesine and cosine functions of said earth's magnetic field direction. 6.The combination set forth in claim 5 wherein said first and secondsignals are supplied to said input switching means are of oppositepolarities.
 7. The combination set forth in claim 5 wherein saidswitching means comprises a first pair of electronic switches, one forconnecting said first signal to said amplifier input and the other forconnecting the amplifier output to the means for modifying said secondsignal and a second pair of electronic switches, one for connecting saidsecond signal to said amplifier input and the other for connecting saidamplifier output to the means for modifying said first signal.
 8. Thecombination set forth in claim 7 wherein said switch controlling meanscomprises a single phase alternating current source and said one pair ofelectronic switches is responsive to one phase thereof and said otherpair of electronic switches is responsive to the opposite phase thereof.9. The combination set forth in claim 4 wherein said means for modifyingsaid first and second signals in accordance with the said amplifieroutput switching means comprises:a first summing circuit responsive tosaid first signal and modified second signal, and a second summingcircuit responsive to said second signal and said modified first signal.10. The combination set forth in claim 4 wherein said means formodifying said first and second signals in accordance with saidamplifier output switching means comprises:a plurality of direct currentsignals of predetermined ratios having values dependent upon the angularorientation of said magnetic field detector inductor elements, means forvarying the relative magnitudes of said direct current signals inaccordance with said amplifier output, and means for applying saidvaried direct current signals to predetermined ones of said detectorinductor elements whereby to vary said plurality of detector signalssupplied to said signal processing means.